1. Field of the Invention
The present invention relates generally to downlink communication systems operating in the presence of multi-path fading, and more particularly, to a method of path gain estimation in a WCDMA system.
2. Description of the Related Art
In mobile communication, the multi-path phenomena of propagated media results in serious fading when transmitted signals are de-constructively added or interfered by other electromagnetic waves transmitted at the same time. Serious signal fading causes the receivers unable to recover the signals properly.
The efficiency of a mobile communication is highly influenced by the channel condition and the transmission environment. Direct-sequence spread spectrum mobile communication systems, such as IS-95, CDMA2000, and WCDMA communication systems often use a RAKE receiver to cope with the multi-path propagation channel, and to cooperate with the handover (hand-off) mechanism of the communication between the mobile station and the base stations. FIG. 1 shows a block diagram of a conventional RAKE receiver used in a spread spectrum system.
The receiving scheme must first confirm the existence of the transmission paths. The receiver then detects the strong and stable signals within the transmission paths, and assigns the signals to corresponding fingers for demodulation. The receiver estimates and tracks the relative delay time of the corresponding transmission paths by a path searching and delay tracking block 102 as shown in FIG. 1. The delay compensation 108 of each finger in the RAKE receiver compensates the propagation delay of the corresponding path. The signal of each finger is despreaded in a dispreading gain compensation block 104, and rotated by a phase angle corresponding to a negative amount of phase rotation introduced by the channel. Subsequent to rotation, signals of each finger are in phase and can be added together. Finally, the path gains and the rotation angles of each finger are estimated by the path gain estimation algorithm in the coder & gain estimation blocks 106 and RAKE combining & SINR estimation block 110.
Generally speaking, there is at least one designated pilot channel in a downlink channel of a mobile communication system in order to reduce the hardware complexity of mobile terminals. The pilot channel provides a phase reference to the mobile terminals within the service area. The base station assigns and sends a phase reference via the pilot channel when a connection is made, thus the mobile terminal does not have to determine the phase reference independently. This is the hypothesis for the path gain estimation algorithm.
The STTD (Space time transmit diversity) scheme has been proposed in the 3GPP specification for use as a next generation mobile system standard. In the STTD scheme, a transmitting device, such as a base station, comprises at least two antennas located apart from each other in a space diversity arrangement.
A signal as shown in FIG. 2(a) originally comprises four bits b0, b1, b2 and b3 in a block, and the signal is encoded into two mutually orthogonal sequences after STTD encoding. For example, the STTD encoder encodes the block (b0, b1, b2, b3) into a first block (b0, b1, b2, b3) for antenna #0, and a second block (−b2, b3, b0, −b1) for antenna #1 as shown in FIG. 2(a). The two STTD encoded blocks transmit through antenna #0 and antenna #1 respectively at the same time. The WCDMA (Wideband Code Division Multiple Access) system uses QPSK (Quadrature Phase Shift Keying) modulation, where two bits are modulated into one symbol. In FIG. 2(b), there are two complex symbols S1 and S2 in a block, which is identical to the input block in FIG. 2(a), as S1 corresponds to bits b0 and b1 and S2 corresponds to bits b2 and b3. FIG. 2(b) shows the STTD encoded process symbolically, wherein the input block of the STTD encoder is (S1, S2), and the output blocks are (S1, S2) and (−S2*, S1*). The notation * represents the complex conjugate of a complex number. If a channel is STTD encoded, the primary common pilot channel (P-CPICH) is specified as the phase reference. The two mutual orthogonal sequences transmitted by two different antennas will both be picked up by the corresponding mobile terminal.
Furthermore, if the complex path gains of antenna #0 and antenna #1 are assumed to be h0, and h1 respectively, the following equations may be derived:
                              r          0                =                                            s              0                        ⁢                          h              0                                -                                    s              1              *                        ⁢                          h              1                                                          (        1        )                                                      r            1                    =                                                                      s                  1                                ⁢                                  h                  0                                            +                                                s                  0                  *                                ⁢                                                      h                    1                                    ⁢                                                                          (                                                                                                              r                          0                                                                                                                                                              r                          1                                                                                                      )                                                      =                                          (                                                                                                    s                        0                                                                                                            -                                                  s                          1                          *                                                                                                                                                                        s                        1                                                                                                            s                        0                        *                                                                                            )                            ⁢                              (                                                                                                    h                        0                                                                                                                                                h                        1                                                                                            )                                                    ⁢                                  ⁢                              (                                                                                r                    0                    *                                                                                                                    r                    1                                                                        )                    =                                    (                                                                                          h                      0                      *                                                                                                  -                                              h                        1                        *                                                                                                                                                        h                      1                                                                                                  h                      0                                                                                  )                        ⁢                          (                                                                                          s                      0                      *                                                                                                                                  s                      1                                                                                  )                                                          (        2        )            
The complex symbols S0 and S1 transmitted in the pilot sequence are known by both the transmitter and the receivers before transmission. The received values r0 and r1 are obtained by despreading the signal transmitted in the pilot channel. The path gains h0 and h1 are thus estimated by solving the simultaneous equations (1) and (2).
Once the path gains h0 and h1 are obtained from the information sent by the pilot channel, simultaneous equations for estimating symbols carried in the data channel can be derived. The symbols transmitted in the data channel can be obtained by substituting the corresponding received values r0 and r1 into the simultaneous equations. The RAKE receiver then combines the symbols acquired from each receiving path, and outputs the combined data to a channel decoder.
The arrangement of transmitted bits in the primary-CPICH (P-CPICH) and secondary-CPICH(S-CPICH) of the WCDMA system is shown in FIG. 4. This design treats the two QPSK symbols s0 and s1 in a data block as the same symbol, i.e. s0=s1, thus further simplifying the path gain estimation calculation.
As shown in FIG. 3(a), each radio frame contains 15 time slots, and each time slot carries 10 symbols. FIG. 3(b) illustrates the modulation pattern of the symbols transmitted through antenna 1 and antenna 2, where A=1+j. As shown in FIG. 5, a block type STTD path gain estimation process first divides the data blocks into two different block types. Each data block contains two QPSK symbols, equivalent to 4 bits of data. The patterns of block type #0 and block type #1 are shown in FIG. 5, and these two block types are transmitted alternatively. The example in FIG. 5 is identical to the example in FIGS. 3 and 4 as symbol A in FIG. 3 represents binary bits (00) and symbol −A represents binary bits (11).
According to the gray-encoding rule, the bit pair (11) corresponds to a QPSK symbol −s0 if the bit pair (00) corresponds to a QPSK symbol S0. Furthermore, if the path gains of the two antennas are assumed to be constant during the transmission of each data block, the relationship between the received values (r0, r1) and the path gains (h0, h1) of the two antennas within the time interval of transmitting block type #0 is expressed in Equations (3) and (4).r0=h0·s0+h1·s0  (3)r1=h0·s0−h1·s0  (4)
Similarly, the relationship between the received values (r0, r1) and the path gains (h0 , h1) of the two antennas within the time interval of transmitting block type #1 is expressed in Equations (5) and (6).r0=h0·s0−h1·s0  (5)r1=h0·s0=h1·s0  (6)
According to the above equations (3), (4), (5) and (6), the path gains h0 and h1 of antenna #0 and antenna #1 respectively can be determined after receiving the pilot symbol. Then the STTD encoded data transmitted in the data channel can hence be decoded according to equations (1) and (2).
In order to reduce the noise of the transmitted signals, each of the pass gain sequences h0 and h1 of the two antennas passes into a low pass filter respectively. The cutoff frequency of the low pass filters is higher than the sum of the maximum carrier frequency offset and the maximum Doppler frequency.
The previously described path gain estimation method however has several shortcomings when implemented in mobile terminal.
The path gain estimation method of the related art is not suitable for mobile terminals as they are not constant, but vary with time due to rapid movement, thus the assumption of constant path gains in the related art introduces enormous errors.
The path gain estimation method of the related art is unfavorable when a mismatch occurs between the carrier frequency of the transmitter (ie. base station) and the carrier frequency of the receiver (ie. mobile terminal). The block type STTD path gain estimation method produces serious jitters when the carrier frequency is shifted beyond an acceptable level (for example, 0.1 ppm).
Another drawback of the related art is the high cost of hardware implementation, since low pass filters are required in the path gain estimation to reduce noise.
As a result of the shortcomings found in the related art, an improved method of time-variant path gain estimation and a system thereof are provided in the present invention.